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1 lt1373 250khz low supply current high efficiency 1.5a switching regulator n boost regulators n ccfl backlight driver n laptop computer supplies n multiple output flyback supplies n inverting supplies the lt ? 1373 is a low supply current high frequency current mode switching regulator. it can be operated in all standard switching configurations including boost, buck, flyback, forward, inverting and cuk. a 1.5a high effi- ciency switch is included on the die, along with all oscilla- tor, control and protection circuitry. all functions of the lt1373 are integrated into 8-pin so/pdip packages. compared to the 500khz lt1372, which draws 4ma of quiescent current, the lt1373 switches at 250khz, typi- cally consumes only 1ma and has higher efficiency. high frequency switching allows for small inductors to be used. all surface mount components consume less than 0.6 square inch of board space. new design techniques increase flexibility and maintain ease of use. switching is easily synchronized to an exter- nal logic level source. a logic low on the shutdown pin reduces supply current to 12 m a. unique error amplifier circuitry can regulate positive or negative output voltage while maintaining simple frequency compensation tech- niques. nonlinear error amplifier transconductance re- duces output overshoot on start-up or overload recovery. oscillator frequency shifting protects external compo- nents during overload conditions. n 1ma i q at 250khz n uses small inductors: 15 m h n all surface mount components n only 0.6 square inch of board space n low minimum supply voltage: 2.7v n constant frequency current mode n current limited power switch: 1.5a n regulates positive or negative outputs n shutdown supply current: 12 m a typ n easy external synchronization n 8-pin so or pdip packages , ltc and lt are registered trademarks of linear technology corporation. output current (ma) 1 70 efficiency (%) 80 90 10 100 1000 lt1373 ?ta02 60 50 100 v in = 5v f = 250khz lt1373 v in v c 5v 1 2 8 5 4 6, 7 sumida cd75-220kc (22 m h) or coilcraft d03316-153 (15 m h) avx tpsd226m025r0200 gnd fb lt1373 ?ta01 v sw s/s l1* 22 m h c1** 22 m f c4** 22 m f c2 0.01 m f r3 5k r2 24.9k 1% r1 215k 1% * ** v out ? 12v ? max i out d1 mbrs120t3 on off l1 15 m h 22 m h i out 0.3a 0.35a + + 5v-to-12v boost converter 12v output efficiency features descriptio u applicatio s u typical applicatio u
2 lt1373 consult factory for military grade parts. (note 1) supply voltage ....................................................... 30v switch voltage lt1373 ............................................................... 35v lt1373hv .......................................................... 42v s/s pin voltage ....................................................... 30v feedback pin voltage (transient, 10ms) .............. 10v feedback pin current ........................................... 10ma negative feedback pin voltage (transient, 10ms) ............................................. 10v operating junction temperature range commercial ........................................ 0 c to 125 c* industrial ......................................... C 40 c to 125 c short circuit ......................................... 0 c to 150 c storage temperature range ................ C 65 c to 150 c lead temperature (soldering, 10 sec)................. 300 c lt1373cn8 lt1373hvcn8 lt1373cs8 lt1373hvcs8 lt1373in8 lt1373hvin8 lt1373is8 lt1373hvis8 s8 part marking order part number 1373h 1373hi 1373 1373i *units shipped prior to date code 9552 are rated at 100 c maximum operating temperature. symbol parameter conditions min typ max units v ref reference voltage measured at feedback pin 1.230 1.245 1.260 v v c = 0.8v l 1.225 1.245 1.265 v i fb feedback input current v fb = v ref 50 150 na l 275 na reference voltage line regulation 2.7v v in 25v, v c = 0.8v l 0.01 0.03 %/v v nfb negative feedback reference voltage measured at negative feedback pin C 2.51 C 2.45 C 2.39 v feedback pin open, v c = 0.8v l C 2.55 C 2.45 C 2.35 v i nfb negative feedback input current v nfb = v nfr l C12 C7 C2 m a negative feedback reference voltage 2.7v v in 25v, v c = 0.8v l 0.01 0.05 %/v line regulation g m error amplifier transconductance d i c = 5 m a 250 375 500 m mho l 150 600 m mho error amplifier source current v fb = v ref C 150mv, v c = 1.5v l 25 50 90 m a error amplifier sink current v fb = v ref + 150mv, v c = 1.5v l 850 1500 m a error amplifier clamp voltage high clamp, v fb = 1v 1.70 1.95 2.30 v low clamp, v fb = 1.5v 0.25 0.40 0.52 v a v error amplifier voltage gain 250 v/ v v c pin threshold duty cycle = 0% 0.8 1 1.25 v f switching frequency 2.7v v in 25v 225 250 275 khz 0 c t j 125 c l 210 250 290 khz C40 c t j 0 c (i grade) 200 290 khz maximum switch duty cycle l 85 95 % switch current limit blanking time 340 500 ns the l denotes specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25 c. v in = 5v, v c = 0.6v, v fb = v ref , v sw , s/s and nfb pins open, unless otherwise noted. 1 2 3 4 8 7 6 5 top view v c fb nfb s/s v sw gnd gnd s v in n8 package 8-lead pdip s8 package 8-lead plastic so t jmax = 125 c, q ja = 100 c/ w (n8) t jmax = 125 c, q ja = 120 c/ w (s8) absolute axi u rati gs w ww u package/order i for atio uu w electrical characteristics
3 lt1373 symbol parameter conditions min typ max units bv output switch breakdown voltage lt1373 l 35 47 v lt1373hv 0 c t j 125 c l 42 47 v C 40 c t j 0 c (i grade) 40 v v sat output switch on resistance i sw = 1a l 0.5 0.85 w i lim switch current limit duty cycle = 50% l 1.5 1.9 2.7 a duty cycle = 80% (note 2) l 1.3 1.7 2.5 a d i in supply current increase during switch on-time 10 20 ma/a d i sw control voltage to switch current 2a/v transconductance minimum input voltage l 2.4 2.7 v i q supply current 2.7v v in 25v l 1 1.5 ma shutdown supply current 2.7v v in 25v, v s/s 0.6v 0 c t j 125 c l 12 30 m a C 40 c t j 0 c (i grade) 50 m a shutdown threshold 2.7v v in 25v l 0.6 1.3 2 v shutdown delay l 5 12 100 m s s/s pin input current 0v v s/s 5v l C10 15 m a synchronization frequency range l 300 340 khz note 1: absolute maximum ratings are those values beyond which the life of the device may be impaired. switch current (a) 0 switch saturation voltage (v) 0.6 0.8 1.0 1.6 lt1373 ?g01 0.4 0.2 0.5 0.7 0.9 0.3 0.1 0 0.4 0.8 1.2 2.0 1.4 0.2 0.6 1.0 1.8 100 c 150 c 25 c ?5 c switch saturation voltage vs switch current temperature ( c) ?0 1.8 input voltage (v) 2.0 2.2 2.4 2.6 050 100 150 lt1373 ?g03 2.8 3.0 ?5 25 75 125 minimum input voltage vs temperature duty cycle (%) 0 switch current limit (a) 1.0 2.0 3.0 0.5 1.5 2.5 20 40 60 80 lt1373 ?g02 100 10 0 30 50 70 90 25 c and 125 c ?5 c switch current limit vs duty cycle the l denotes specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25 c. v in = 5v, v c = 0.6v, v fb = v ref , v sw , s/s and nfb pins open, unless otherwise noted. electrical characteristics note 2: for duty cycles (dc) between 50% and 90%, minimum guaranteed switch current is given by i lim = 0.667 (2.75 C dc). typical perfor a ce characteristics uw
4 lt1373 shutdown delay and threshold vs temperature error amplifier output current vs feedback pin voltage temperature ( c) ?0 0 shutdown delay ( m s) shutdown threshold (v) 2 6 8 10 20 14 0 50 75 lt1373 ?g04 4 16 18 12 0 0.2 0.6 0.8 1.0 2.0 1.4 0.4 1.6 1.8 1.2 ?5 25 100 125 150 shutdown threshold shutdown delay temperature ( c) ?0 0 minimum synchronization voltage (v p-p ) 0.5 1.0 1.5 2.0 050 100 150 lt1373 ?g05 2.5 3.0 ?5 25 75 125 f sync = 330khz minimum synchronization voltage vs temperature feedback pin voltage (v) 100 error amplifier output current ( m a) ?5 ?0 ?5 75 25 0.1 0.1 50 0 0.3 0.2 v ref ?5 c 125 c 25 c lt1373 ?g06 s/s pin input current vs voltage s/s pin voltage (v) ? s/s pin input current ( m a) 1 3 5 7 lt1373 ?g07 ? ? 0 2 4 ? ? ? 1 3 5 08 2 4 6 9 v in = 5v error amplifier transconductance vs temperature switching frequency vs feedback pin voltage feedback pin voltage (v) 0 switching frequency (% of typical) 70 90 110 0.8 lt1373 ?g08 50 30 60 80 100 40 20 10 0.2 0.4 0.6 0.1 0.9 0.3 0.5 0.7 1.0 temperature ( c) ?0 0 transconductance ( m mho) 200 500 0 50 75 lt1373 ?g09 100 400 300 ?5 25 100 125 150 g m = d i (v c ) d v (fb) v c pin threshold and high clamp voltage vs temperature negative feedback input current vs temperature feedback input current vs temperature temperature ( c) ?0 0.4 v c pin voltage (v) 0.6 1.0 1.2 1.4 2.4 1.8 0 50 75 lt1373 ?g10 0.8 2.0 2.2 1.6 ?5 25 100 125 150 v c high clamp v c threshold temperature ( c) ?0 feedback input current (na) 200 250 300 150 lt1373 ?g11 150 100 0 0 50 100 50 400 350 ?5 25 75 125 v fb = v ref temperature ( c) ?0 ?0 negative feedback input current ( m a) ?2 ?4 0 0 50 75 lt1373 ?g12 ?6 ?8 ? ? ? ? ?0 ?5 25 100 125 150 v nfb = v nfr typical perfor a ce characteristics uw
5 lt1373 v c (pin 1): compensation pin. the v c pin is used for frequency compensation, current limiting and soft start. it is the output of the error amplifier and the input of the current comparator. loop frequency compensation can be performed with an rc network connected from the v c pin to ground. fb (pin 2): t he feedback pin is used for positive output voltage sensing and oscillator frequency shifting. it is the inverting input to the error amplifier. the noninverting input of this amplifier is internally tied to a 1.245v reference. load on the fb pin should not exceed 100 m a when the nfb pin is used. see applications information. nfb (pin 3): the negative feedback pin is used for negative output voltage sensing. it is connected to the inverting input of the negative feedback amplifier through a 400k source resistor. s/s (pin 4): shutdown and synchronization pin. the s/s pin is logic level compatible. shutdown is active low and the shutdown threshold is typically 1.3v. for normal operation, pull the s/s pin high, tie it to v in or leave it floating. to synchronize switching, drive the s/s pin be- tween 300khz and 340khz. v in (pin 5): input supply pin. bypass v in with 10 m f or more. the part goes into undervoltage lockout when v in drops below 2.5v. undervoltage lockout stops switching and pulls the v c pin low. gnd s (pin 6): the ground sense pin is a clean ground. the internal reference, error amplifier and negative feed- back amplifier are referred to the ground sense pin. con- nect it to ground. keep the ground path connection to the output resistor divider and the v c compensation network free of large ground currents. gnd (pin 7): the ground pin is the emitter connection of the power switch and has large currents flowing through it. it should be connected directly to a good quality ground plane. v sw (pin 8): the switch pin is the collector of the power switch and has large currents flowing through it. keep the traces to the switching components as short as possible to minimize radiation and voltage spikes. + negative feedback amp nfb s/s fb 400k 200k 0.08 w + v c v in gnd lt1373 ?bd gnd sense 1.245v ref 5:1 frequency shift 250khz osc sync shutdown delay and reset low dropout 2.3v reg anti-sat logic driver sw switch + a v ? 6 comp error amp current amp uu u pi fu ctio s block diagra w
6 lt1373 the lt1373 is a current mode switcher. this means that switch duty cycle is directly controlled by switch current rather than by output voltage. referring to the block diagram, the switch is turned on at the start of each oscillator cycle. it is turned off when switch current reaches a predetermined level. control of output voltage is obtained by using the output of a voltage sensing error amplifier to set current trip level. this technique has several advantages. first, it has immediate response to input voltage variations, unlike voltage mode switchers which have notoriously poor line transient response. second, it reduces the 90 phase shift at mid-frequencies in the energy storage inductor. this greatly simplifies closed-loop frequency compensation under widely vary- ing input voltage or output load conditions. finally, it allows simple pulse-by-pulse current limiting to provide maximum switch protection under output overload or short conditions. a low dropout internal regulator pro- vides a 2.3v supply for all internal circuitry. this low dropout design allows input voltage to vary from 2.7v to 25v with virtually no change in device performance. a 250khz oscillator is the basic clock for all internal timing. it turns on the output switch via the logic and driver circuitry. special adaptive anti-sat circuitry detects onset of saturation in the power switch and adjusts driver current instantaneously to limit switch saturation. this minimizes driver dissipation and provides very rapid turn-off of the switch. a 1.245v bandgap reference biases the positive input of the error amplifier. the negative input of the amplifier is brought out for positive output voltage sensing. the error amplifier has nonlinear transconductance to reduce out- put overshoot on start-up or overload recovery. when the feedback voltage exceeds the reference by 40mv, error amplifier transconductance increases ten times, which reduces output overshoot. the feedback input also invokes oscillator frequency shifting, which helps pro- tect components during overload conditions. when the feedback voltage drops below 0.6v, the oscillator fre- quency is reduced 5:1. lower switching frequency allows full control of switch current limit by reducing minimum switch duty cycle. unique error amplifier circuitry allows the lt1373 to directly regulate negative output voltages. the negative feedback amplifiers 400k source resistor is brought out for negative output voltage sensing. the nfb pin regulates at C 2.45v while the amplifier output internally drives the fb pin to 1.245v. this architecture, which uses the same main error amplifier, prevents duplicating functions and maintains ease of use. (consult linear technology mar- keting for units that can regulate down to C 1.25v.) the error signal developed at the amplifier output is brought out externally. this pin (v c ) has three different functions. it is used for frequency compensation, current limit adjustment and soft starting. during normal regula- tor operation this pin sits at a voltage between 1v (low output current) and 1.9v (high output current). the error amplifier is a current output (g m ) type, so this voltage can be externally clamped for lowering current limit. like- wise, a capacitor coupled external clamp will provide soft start. switch duty cycle goes to zero if the v c pin is pulled below the control pin threshold, placing the lt1373 in an idle mode. positive output voltage setting the lt1373 develops a 1.245v reference (v ref ) from the fb pin to ground. output voltage is set by connecting the fb pin to an output resistor divider (figure 1). the fb pin bias current represents a small error and can usually be ignored for values of r2 up to 25k. the suggested value for r2 is 24.9k. the nfb pin is normally left open for positive output applications. r1 v out = v ref 1 + r2 fb pin v ref v out () r1 r2 r1 = r2 ?1 () v out 1.245 lt1373 ?f01 figure 1. positive output resistor divider operatio u applicatio s i for atio wu uu
7 lt1373 negative output voltage setting the lt1373 develops a C 2.45v reference (v nfr ) from the nfb pin to ground. output voltage is set by connecting the nfb pin to an output resistor divider (figure 2). the C 7 m a nfb pin bias current (i nfb ) can cause output voltage errors and should not be ignored. this has been accounted for in the formula in figure 2. the suggested value for r2 is 2.49k. the fb pin is normally left open for negative output applications. see dual polarity output voltage sensing for limitations of fb pin loading when using the nfb pin. a logic low on the s/s pin activates shutdown, reducing the parts supply current to 12 m a. typical synchronization range is from 1.05 and 1.8 times the parts natural switch- ing frequency, but is only guaranteed between 300khz and 340khz. a 12 m s resetable shutdown delay network guar- antees the part will not go into shutdown while receiving a synchronization signal. caution should be used when synchronizing above 330khz because at higher sync frequencies the ampli- tude of the internal slope compensation used to prevent subharmonic switching is reduced. this type of subharmonic switching only occurs when the duty cycle of the switch is above 50%. higher inductor values will tend to eliminate problems. thermal considerations care should be taken to ensure that the worst-case input voltage and load current conditions do not cause exces- sive die temperatures. the packages are rated at 120 c/w for so (s8) and 130 c/w for pdip (n8). average supply current (including driver current) is: i in = 1ma + dc (i sw /60 + i sw ? 0.004) i sw = switch current dc = switch duty cycle switch power dissipation is given by: p sw = (i sw ) 2 ? r sw ? dc r sw = output switch on resistance total power dissipation of the die is the sum of supply current times supply voltage plus switch power: p d(total) = (i in ? v in ) + p sw choosing the inductor for most applications the inductor will fall in the range of 10 m h to 50 m h. lower values are chosen to reduce physical size of the inductor. higher values allow more output current because they reduce peak current seen by the power switch which has a 1.5a limit. higher values also reduce input ripple voltage, and reduce core loss. when choosing an inductor you might have to consider maximum load current, core and copper losses, allowable r1 ? out = v nfb + i nfb (r1) 1 + r2 lt1373 ?f02 nfb pin v nfr i nfb ? out () r1 r2 r1 = + (7 ?10 ? ) ? v out ? ?2.45 () 2.45 r2 figure 2. negative output resistor divider dual polarity output voltage sensing certain applications benefit from sensing both positive and negative output voltages. one example is the dual output flyback converter with overvoltage protection circuit shown in the typical applications section. each output voltage resistor divider is individually set as de- scribed above. when both the fb and nfb pins are used, the lt1373 acts to prevent either output from going beyond its set output voltage. for example in this applica- tion, if the positive output were more heavily loaded than the negative, the negative output would be greater and would regulate at the desired set-point voltage. the posi- tive output would sag slightly below its set-point voltage. this technique prevents either output from going unregu- lated high at no load. please note that the load on the fb pin should not exceed 100 m a when the nfb pin is used. this situation occurs when the resistor dividers are used at both fb and nfb. true load on fb is not the full divider current unless the positive output is shorted to ground. see dual output flyback converter application. shutdown and synchronization the dual function s/s pin provides easy shutdown and synchronization. it is logic level compatible and can be pulled high, tied to v in or left floating for normal operation. applicatio s i for atio wu uu
8 lt1373 component height, output voltage ripple, emi, fault cur- rent in the inductor, saturation, and of course, cost. the following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements. 1. assume that the average inductor current (for a boost converter) is equal to load current times v out /v in and decide whether or not the inductor must withstand continuous overload conditions. if average inductor current at maximum load current is 0.5a, for instance, a 0.5a inductor may not survive a continuous 1.5a overload condition. also, be aware that boost convert- ers are not short-circuit protected, and that under output short conditions, inductor current is limited only by the available current of the input supply. 2. calculate peak inductor current at full load current to ensure that the inductor will not saturate. peak current can be significantly higher than output current, espe- cially with smaller inductors and lighter loads, so dont omit this step. powered iron cores are forgiving be- cause they saturate softly, whereas ferrite cores satu- rate abruptly. other core materials fall in between somewhere. the following formula assumes continu- ous mode operation, but it errors only slightly on the high side for discontinuous mode, so it can be used for all conditions. i peak = i out v in = minimum input voltage f = 250khz switching frequency + v out v in v in (v out v in ) 2(f)(l)(v out ) 3. decide if the design can tolerate an open core geom- etry like a rod or barrel, which have high magnetic field radiation, or whether it needs a closed core like a toroid to prevent emi problems. one would not want an open core next to a magnetic storage media for instance! this is a tough decision because the rods or barrels are temptingly cheap and small, and there are no helpful guidelines to calculate when the magnetic field radia- tion will be a problem. 4. start shopping for an inductor which meets the require- ments of core shape, peak current (to avoid saturation), average current (to limit heating), and fault current, (if the inductor gets too hot, wire insulation will melt and cause turn-to-turn shorts). keep in mind that all good things like high efficiency, low profile and high temperature operation will increase cost, sometimes dramatically. 5. after making an initial choice, consider the secondary things like output voltage ripple, second sourcing, etc. use the experts in the linear technology application department if you feel uncertain about the final choice. they have experience with a wide range of inductor types and can tell you about the latest developments in low profile, surface mounting, etc. output capacitor the output capacitor is normally chosen by its effective series resistance (esr), because this is what determines output ripple voltage. at 500khz, any polarized capacitor is essentially resistive. to get low esr takes volume , so physically smaller capacitors have high esr. the esr range for typical lt1373 applications is 0.05 w to 0.5 w . a typical output capacitor is an avx type tps, 22 m f at 25v, with a guaranteed esr less than 0.2 w . this is a d size surface mount solid tantalum capacitor. tps capacitors are specially constructed and tested for low esr, so they give the lowest esr for a given volume. to further reduce esr, multiple output capacitors can be used in parallel. the value in microfarads is not particularly critical and values from 22 m f to greater than 500 m f work well, but you cannot cheat mother nature on esr. if you find a tiny 22 m f solid tantalum capacitor, it will have high esr and output ripple voltage will be terrible. table 1 shows some typical solid tantalum surface mount capacitors. table 1. surface mount solid tantalum capacitor esr and ripple current e case size esr (max w ) ripple current (a) avx tps, sprague 593d 0.1 to 0.3 0.7 to 1.1 avx taj 0.7 to 0.9 0.4 d case size avx tps, sprague 593d 0.1 to 0.3 0.7 to 1.1 avx taj 0.9 to 2.0 0.36 to 0.24 c case size avx tps 0.2 (typ) 0.5 (typ) avx taj 1.8 to 3.0 0.22 to 0.17 b case size avx taj 2.5 to 10 0.16 to 0.08 applicatio s i for atio wu uu
9 lt1373 many engineers have heard that solid tantalum capacitors are prone to failure if they undergo high surge currents. this is historically true and type tps capacitors are specially tested for surge capability, but surge ruggedness is not a critical issue with the output capacitor. solid tantalum capacitors fail during very high turn-on surges, which do not occur at the output of regulators. high discharge surges, such as when the regulator output is dead shorted, do not harm the capacitors. single inductor boost regulators have large rms ripple current in the output capacitor, which must be rated to handle the current. the formula to calculate this is: output capacitor ripple current (rms) i ripple (rms) = i out = i out v out v in v in dc 1 ?dc input capacitors the input capacitor of a boost converter is less critical due to the fact that the input current waveform is triangular, and does not contain large squarewave currents as is found in the output capacitor. capacitors in the range of 10 m f to 100 m f with an esr (effective series resistance) of 0.3 w or less work well up to a full 1.5a switch current. higher esr capacitors may be acceptable at low switch currents. input capacitor ripple current for boost con- verter is: i ripple = 0.3(v in )(v out ?v in ) (f)(l)(v out ) f = 250khz switching frequency the input capacitor can see a very high surge current when a battery or high capacitance source is connected live, and solid tantalum capacitors can fail under this condition. several manufacturers have developed a line of solid tantalum capacitors specially tested for surge capability (avx tps series, for instance), but even these units may fail if the input voltage approaches the maximum voltage rating of the capacitor. avx recommends derating capaci- tor voltage by 2:1 for high surge applications. ceramic and aluminum electrolytic capacitors may also be used and have a high tolerance to turn-on surges. ceramic capacitors higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. these are tempt- ing for switching regulator use because of their very low esr. unfortunately, the esr is so low that it can cause loop stability problems. solid tantalum capacitor esr generates a loop zero at 5khz to 50khz that is instrumen- tal in giving acceptable loop phase margin. ceramic ca- pacitors remain capacitive to beyond 300khz and usually resonate with their esl before esr becomes effective. they are appropriate for input bypassing because of their high ripple current ratings and tolerance of turn-on surges. linear technology plans to issue a design note on the use of ceramic capacitors in the near future. output diode the suggested output diode (d1) is a 1n5818 schottky or its motorola equivalent, mbr130. it is rated at 1a average forward current and 30v reverse voltage. typical forward voltage is 0.42v at 1a. the diode conducts current only during switch-off time. peak reverse voltage for boost converters is equal to regulator output voltage. average forward current in normal operation is equal to output current. frequency compensation loop frequency compensation is performed on the output of the error amplifier (v c pin) with a series r c network. the main pole is formed by the series capacitor and the output impedance ( ? 1m w ) of the error amplifier. the pole falls in the range of 5hz to 30hz. the series resistor creates a zero at 2khz to 10khz, which improves loop stability and transient response. a second capacitor, typically one tenth the size of the main compensation capacitor, is sometimes used to reduce the switching frequency ripple on the v c pin. v c pin ripple is caused by output voltage ripple attenuated by the output divider and multiplied by the error amplifier. without the second capacitor, v c pin ripple is: v c pin ripple = 1.245(v ripple )(g m )(r c ) v out applicatio s i for atio wu uu
10 lt1373 v ripple = output ripple (v p-p ) g m = error amplifier transconductance ( ? 375 m mho) r c = series resistor on v c pin v out = dc output voltage to prevent irregular switching, v c pin ripple should be kept below 50mv p-p . worst-case v c pin ripple occurs at maximum output load current and will also be increased if poor quality (high esr) output capacitors are used. the addition of a 0.001 m f capacitor on the v c pin reduces switching frequency ripple to only a few millivolts. a low value for r c will also reduce v c pin ripple, but loop phase margin may be inadequate. switch node considerations for maximum efficiency, switch rise and fall time are made as short as possible. to prevent radiation and high fre- quency resonance problems, proper layout of the compo- nents connected to the switch node is essential. b field (magnetic) radiation is minimized by keeping output di- ode, switch pin and output bypass capacitor leads as short as possible. e field radiation is kept low by minimizing the length and area of all traces connected to the switch pin. a ground plane should always be used under the switcher circuitry to prevent interplane coupling. positive-to-negative converter with direct feedback lt1373 v in v c v in 2.7v to 16v 1 3 ? max i out 8 5 4 6, 7 *coiltronics ctx20-2 (407) 241-7876 gnd nfb lt1373 ?ta03 v sw s/s d2 p6ke-15a d3 1n4148 d1 mbrs130lt3 c1 22 m f c2 0.01 m f r1 5k r3 2.49k 1% r2 2.55k 1% ? out ? ?v c3 47 m f on off v in 3v 5v 9v i out 0.3a 0.5a 0.75a 2 1 4 t1* 3 + + dual output flyback converter with overvoltage protection lt1373 v in fb v c v in 4.75v to 13v 1 3 8 5 2 4 6, 7 *dale lpe-4841-100mb (605) 665-9301 gnd nfb lt1373 ?ta04 v sw s/s p6ke-20a 1n4148 mbrs140t3 mbrs140t3 c1 100 m f r2 275k 1% r1 302.6k 1% c2 0.01 m f r3 5k r5 2.49k 1% r4 12.4k 1% ? out ?5v v out 15v c3 47 m f c4 47 m f on off 2, 3 6, 7 5 t1* 4 8 1 + + + typical applicatio n s n u the high speed switching current path is shown schemati- cally in figure 3. minimum lead length in this path is essential to ensure clean switching and low emi. the path including the switch, output diode and output capacitor is the only one containing nanosecond rise and fall times. keep this path as short as possible. more help for more detailed information on switching regulator circuits, please see an19. linear technology also offers a computer software program, switchercad tm , to assist in designing switching converters. in addition, our applica- tions department is always ready to lend a helping hand. load v out l1 switch node lt1373 ?f03 v in high frequency circulating path figure 3 applicatio s i for atio wu uu switchercad is a trademark of linear technology corporation.
11 lt1373 low ripple 5v to C 3v cuk ? converter information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of circuits as described herein will not infringe on existing patent rights. lt1373 v in s/s gnd gnd s v sw nfb v c 5 4 7 6 8 3 1 + + r4 5k r2 5.49k 1% r1 1k 1% c4 0.01 m f c6 0.1 m f v out ?v 250ma lt1373 ?ta05 v in 5v c3 47 m f 16v c1 22 m f 10v c2 47 m f 16v 4 1 3 l1* 2 d1** sumida cls62-100l motorola mbr0520lt3 patents may apply * ** ? + typical applicatio s u package descriptio n u dimensions in inches (millimeters) unless otherwise noted. n8 package 8-lead pdip (narrow 0.300) (ltc dwg # 05-08-1510) s8 package 8-lead plastic small outline (narrow 0.150) (ltc dwg # 05-08-1610) n8 1098 0.009 ?0.015 (0.229 ?0.381) 0.300 ?0.325 (7.620 ?8.255) 0.325 +0.035 0.015 +0.889 0.381 8.255 () 0.100 (2.54) bsc 0.065 (1.651) typ 0.045 ?0.065 (1.143 ?1.651) 0.130 0.005 (3.302 0.127) 0.020 (0.508) min 0.018 0.003 (0.457 0.076) 0.125 (3.175) min 12 3 4 87 6 5 0.255 0.015* (6.477 0.381) 0.400* (10.160) max *these dimensions do not include mold flash or protrusions. mold flash or protrusions shall not exceed 0.010 inch (0.254mm) 0.016 ?0.050 (0.406 ?1.270) 0.010 ?0.020 (0.254 ?0.508) 45 0 ?8 typ 0.008 ?0.010 (0.203 ?0.254) so8 1298 0.053 ?0.069 (1.346 ?1.752) 0.014 ?0.019 (0.355 ?0.483) typ 0.004 ?0.010 (0.101 ?0.254) 0.050 (1.270) bsc 1 2 3 4 0.150 ?0.157** (3.810 ?3.988) 8 7 6 5 0.189 ?0.197* (4.801 ?5.004) 0.228 ?0.244 (5.791 ?6.197) dimension does not include mold flash. mold flash shall not exceed 0.006" (0.152mm) per side dimension does not include interlead flash. interlead flash shall not exceed 0.010" (0.254mm) per side * **
12 lt1373 ? linear technology corporation 1995 sn1373 1373fbs lt/tp 0200 2k rev b ? printed in the usa related parts part number description comments lt1172 100khz 1.25a boost switching regulator also for flyback, buck and inverting configurations ltc ? 1265 13v 1.2a monolithic buck converter includes pmos switch on-chip lt1302 micropower 2a boost converter converts 2v to 5v at 600ma lt1308a/lt1308b 600khz 2a switch dc/dc converter 5v at 1a from a single li-ion cell lt1370 500khz high efficiency 6a boost converter 6a, 0.065 w internal switch lt1372 500khz 1.5a boost switching regulator also regulates negative flyback outputs lt1374 4.5a, 500khz step-down converter 4.5a, 0.07 w internal switch lt1376 500khz 1.5a buck switching regulator handles up to 25v inputs lt1377 1mhz 1.5a boost switching regulator only 1mhz integrated switching regulator available lt1613 1.4mhz switching regulator in 5-lead sot-23 5v at 200ma from 4.4v input lt1615 micropower step-up dc/dc in 5-lead sot-23 20 m a i q , 36v, 350ma switch lt1949 600khz, 1a switch pwm dc/dc converter 1.1a, 0.5 w , 30v internal switch, v in as low as 1.5v d2 1n4148 q2 1n5818 d1 1n4148 562 w * 20k dimming 10k 330 w 10 1 2 3 4 5 q1 10 m f c1 0.1 m f v in 4.5v to 30v v in v sw v fb v c gnd s/s 5 8 4 2 1 6, 7 lt1373 2 m f 0.1 m f l1 100 m h t1 lt1372 ?ta06 c1 = wima mkp-20 l1 = coilcraft d03316-104 t1 = coiltronics ctx 110609 * = 1% film resistor do not substitute components q1, q2 = zetex ztx849 or rohm 2sc5001 lamp c2 27pf 5ma max 2.2 m f 2.7v to 5.5v 22k 1n4148 optional remote dimming coiltronics (407) 241-7876 coilcraft (708) 639-6400 on off ccfl backlight application circuits contained in this data sheet are covered by u.s. patent number 5408162 and other patents pending + + + 90% efficient ccfl supply lt1373 v in v c v in 4v to 9v 1 2 ? max i out c1 = avx tpsd 336m020r0200 c2 = tokin 1e225zy5u-c203-f c3 = avx tpsd 107m010r0100 l1 = coiltronics ctx33-2, single inductor with two windings 8 5 4 6, 7 gnd fb lt1373 ?ta07 v sw s/s d1 mbrs130lt3 c1 33 m f 20v c4 0.01 m f c2 2.2 m f r1 5k r3 24.9k 1% r2 75k 1% v out ? 5v on off v in 4v 5v 7v 9v i out 0.45a 0.55a 0.65a 0.72a l1a 33 m h l1b 33 m h c3 100 m f 10v + + two li-ion cells to 5v sepic conveter typical applicatio s u linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 l fax: (408) 434-0507 l www.linear-tech.com

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